Method and apparatus to cancel additive sinusoidal disturbances in ofdm receivers

ABSTRACT

Orthogonal frequency division multiplexing (OFDM) has become a popular transmission method for high speed wireless radio transmission, due to its potential for low complexity of transmitters and receivers. A method and apparatus are contemplated for cancelling additive sinusoidal disturbances of a known frequency in OFDM receivers which arise e.g. from clock signals that are present for frequency reference, mixer control, and A/D converter control, as well as harmonics and mixing products of those periodic signals, coupling into some point in the receiver chain and appearing as rotating complex exponentials superimposed to complex baseband receive signals. According to the inventive method and apparatus an estimation of an amplitude and phase of a disturbing superimposed tone with a known frequency is obtained and the amplitude and phase estimation is used to cancel the spurious tone preventing a degradation of receiver sensitivity while achieving low implementation complexity.

The present invention relates to a method and an apparatus to canceladditive sinusoidal disturbances of a known frequency in OFDM receivers.

BACKGROUND OF THE INVENTION

Orthogonal frequency division multiplexing (OFDM) has become a populartransmission method for high-speed wireless radio transmission, due toits potential for low complexity of transmitters and receivers, pairedwith robustness under severe multi-path conditions. A more detaileddiscussion on OFDM in found in S. B. Weinstein and P. M. Ebert: Datatransmission by frequency-division multiplexing using the discreteFourier transform. IEEE Trans. Communication Technology,COM-19(5):628-634, Oct. 1971. The wired counterpart, known as discretemulti-tone (DMT) employs similar techniques. The transmitter uses aninverse discrete Fourier transform (IDFT) to generate a multi-carriersignal, and the receiver applies the Discrete Fourier Transform (DFT) todemodulate the data.

Integrated radio receivers need a large gain and a low noise figure toachieve a high sensitivity. Clock signals which are present forfrequency reference, mixer control, and A/D converter control, as wellas harmonics and mixing products of these periodic signals, may coupleinto some point in the receiver chain and appear as rotating complexexponentials superimposed to the complex baseband receive signal. If thelevel of such tones becomes too high, they may degrade the receiversensitivity. The frequencies of such disturbing tones originating fromthe RF receiver itself are directly related to the clock frequenciesoccurring in the receiver.

As stated above, unwanted tones superimposed to the received signal mayreduce the receiver sensitivity. The safest approach to prevent thisproblem is to directly avoid the occurrence of such tones. Even thecoupling mechanism may be known and a re-spin of the receiver design maybe able to reduce the coupling. However, in highly integrated receiversystems the effort to achieve this can be quite high, possibly requiringdetailed modelling, design modifications and additional verification.

A general object of the present invention, therefore, is to mitigatesuch additive disturbing tones in an OFDM baseband receiver, whileachieving low implementation complexity.

SUMMARY OF THE INVENTION

According to an aspect of the present invention there is provided amethod for cancelling additive sinusoidal disturbances in OFDM receiversas claimed in claim 1. According to a further aspect of the presentinvention there is provided an apparatus as claimed in claim 16. Theinventive method and apparatus obtain an estimation of an amplitude andphase of a disturbing superimposed tone, whose frequency is known, anduse such amplitude and phase estimation values to cancel the tone suchthat receiver sensitivity degradation is avoided.

In accordance with the invention the implementation is made in a way toachieve a low complexity, which translates into low overhead powerconsumption in applying the method.

Additional features and advantages of the present invention will beapparent from the following detailed description of specific embodimentswhich is given by way of example and in which reference will be made tothe accompanying drawings, wherein:

FIG. 1 is a schematic block diagram of an OFDM receiver, in which thepresent invention may be implemented;

FIG. 2 is a block diagram of a typical known OFDM demodulator;

FIG. 3 shows a block diagram of an OFDM baseband receiver according to apreferred embodiment of the present invention;

FIG. 4 is a block diagram illustrating in more detail a preferredembodiment of the spur cancellation unit of FIG. 3 according to thepresent invention; and

FIG. 5 is a block diagram showing another preferred embodiment of thespur cancellation unit of FIG. 3 according to the present invention.

DETAILED DESCRIPTION

FIG. 1 shows a schematic block diagram of an OFDM receiver 1 in whichthe present invention may be implemented. An analog OFDM radio signal isreceived via an antenna 20 and is fed into a radio receiver 30 where itis converted to a digital complex baseband signal. Typically, radioreceiver 30 consists of a low noise amplifier, a mixer which iscontrolled by a local oscillator, a band selection filter, furtheramplifier stages and optionally a second mixer, an analog-to-digitalconverter, and a digital decimation filter. Radio receiver 30 outputs adigital complex baseband signal. This signal is fed into a digital OFDMbaseband demodulator 40, from where demodulated data are output.

FIG. 2 is a block diagram of a typical OFDM demodulator 40 as shown inFIG. 1, as it is known from the prior art. The input signal which is adigital complex baseband signal supplied from radio receiver 30 of FIG.1 is fed to a guard interval removal unit 410 where it is cut intoblocks of samples of a length corresponding to the OFDM symbol period.Then, the guard period of each such sample block is removed, and aDiscrete Fourier Transform (DFT) is performed on each remainder of thesample blocks in a Discrete Fourier Transform unit 420. DFT unit 420outputs data comprised of symbols which are received on respective OFDMsub-carriers in a data equalization unit 430. Optionally, channelestimation for all sub-carriers of interest is performed in a channelestimation unit 425 prior of being fed into data equalization unit 430.After equalization the sub-carrier symbols are fed from dataequalization unit 430 to a symbol demapper 440 which outputs soft bitsto be fed to a decoder. Discrete Fourier Transform in DFT unit 420 istypically implemented as a Fast Fourier Transform (FFT). This kind ofOFDM demodulator is well known in prior art. However, it has thedrawback that it is not robust against sinusoidal disturbances whichtypically occur by coupling of periodic voltages or currents into the RFsignal path. Such sinusoidal disturbances appear at the input of OFDMbaseband receiver 40 as superimposed complex rotating exponentials.Depending on the level and frequency of such disturbances, a largenumber of information symbols may be corrupted. This degrades thesensitivity of the receiver.

FIG. 3 shows a OFDM baseband receiver 40A modified according to theinvention. OFDM baseband receiver 40A is similar to OFDM basebandreceiver 40 of FIG. 2 described above except that it additionallyincludes a spur cancellation unit 500 right behind DFT unit 420. Thefunction of this spur cancellation unit 500 is to estimate both anamplitude and a phase of a superimposed rotating exponential of knownfrequency which is present at the input of DFT unit 420, and toeliminate this disturbance. Such tones of known frequency typicallyoriginate from harmonics and potential mixing products of periodicsignals occurring in the RF front-end. The frequencies of those signalsare in a constant ratio with the frequency of the reference clock whichis normally used for both the RF front-end and the digital basebandreceiver.

Before referring to FIG. 4 which illustrates spur cancellation unit 500in greater detail, some mathematical basis of the functionality of spurcancellation unit 500 will be delineated, first, for the sake ofunderstanding of the operation thereof, as follows. We make thefollowing definitions:

f_(T) is the frequency of the disturbing tone, normalized to thesampling frequency;

N_(DFT) is the length of the discrete Fourier transform in samples;

N_(Guard) is the length of the guard interval in samples;

N_(Sym)=N_(DFT)+N_(Guard) is the number of time-domain samples per OFDMsymbol;

k is the sampling time index;

y(k)=r(k)+z(k) is the complex baseband receive signal input into OFDMdemodulator 40; with

r(k) being the actual receive signal including other disturbances likenoise; and

z(k)=A_(T)·exp(j2π·f_(T)·k+φ_(T)) being the disturbing superimposedcomplex exponential; with

f_(T) being the known frequency, and A_(T) and φ_(T) being amplitude andphase, respectively, of the disturbing complex exponential, which are tobe estimated.

Assuming that 0≦f_(T)<1, the periodic spectrum of a digital signalallows to map any possible tone onto this range.

Transformation of a complex exponential z(k)|0≦k<N_(DFT) via DFT yieldsthe values

${{Z(n)} = {A_{T} \cdot {\exp \left( {j \cdot \phi_{T}} \right)} \cdot {\sum\limits_{k = 0}^{N_{DFT} - 1}{\exp \left( {j\; 2{\pi \cdot k \cdot \left( {f_{T} - \frac{n}{N_{DFT}}} \right)}} \right)}}}},$

wheren denotes the element index in the resulting vector.Rewriting this equation as

Z(n)=[A _(T)·exp(j·φ _(T))/N _(DFT) ]·W(n)

splits it into the amplitude/phase factor (A_(T)·exp(j·φ_(T))/N_(DFT))and the weighting pattern

${{W(n)} = {\frac{1}{N_{DFT}} \cdot {\sum\limits_{k = 0}^{N_{DFT} - 1}{\exp \left( {j\; 2{\pi \cdot k \cdot \left( {f_{T} - \frac{n}{N_{DFT}}} \right)}} \right)}}}},$

which is only determined by the frequency of the disturbing tone (whentreating N_(DFT) as given). We assume that the frequency of thedisturbing tone is known, and we need to estimate the amplitude and thephase of the tone.Now consider the receive signal after DFT, which ideally consists onlyof a superposition of data symbols disturbed by the channel fading andadditive noise. Let y(k)|K·N_(Sym)≦k<K·N_(Sym)+N_(DFT) denote the DFTinput samples of OFDM symbol number K and

Y_(K)(n) denote the associated DFT output vector, with 0≦n<N_(DFT).

If no additive complex exponential is present, we assume that the outputof all OFDM sub-carriers during reception is a zero-mean random process,i.e.,

E{Y _(K)(n)}=0∀K,n.

Furthermore we assume that distinct DFT output symbols are statisticallyindependent, i.e.,

E{Y _(K) ₁ (n ₁)·Y _(K) ₂ (n ₂)}=0∀K ₁ ≠K ₂ νn ₁ ≠n ₂

Three key ideas are applied for estimation of a superimposed disturbingcomplex exponential:

1. The scalar product of the DFT output vector Y_(K)(n) with the patternW(n|f_(T)),

$P_{K} = {\sum\limits_{n = 0}^{N_{DFT} - 1}{{Y_{K}(n)} \cdot {W^{*}(n)}}}$

is the projection of the DFT output vector into the direction of thetone and is an estimate of the amplitude and phase factor(A_(T)·exp(j·φ_(T))/N_(DFT)) multiplied with a phase offset termexp(j2π·f_(T)·K·N_(Sym)) which is the start of phase of the complexexponential at the beginning of OFDM symbol number K. Hence, with theassumptions made the expectation of the above scalar product isE{P_(K)}=A_(T)·(j·φ_(T))/N_(DFT)·exp(j2π·f_(T)·K·N_(Sym)).2. A back-rotation of the scalar product by the start phase yieldsQ_(K)=P_(K)·exp(−j2π·f_(T)·K·N_(Sym)) which is an estimate of theamplitude and phase with E{Q_(K)}=A_(T)·(j·φ_(T))/N_(DFT)3. Averaging of multiple such back-rotated estimates Q_(K) reduces theestimation error.

FIG. 4 illustrates in more detail a first preferred embodiment of spurcancellation unit of FIG. 3 according to the invention for cancelling adisturbing complex exponential in an OFDM receiver employing theprinciples described above.

An offset phasor F_(K)=exp(j2π·f_(T)·N_(Sym)), here with the constantF_(K)=F∀K is input to an offset phasor accumulation unit 510 and iscumulatively multiplied to obtain a sequence of start phasors

$R_{K} = {{\prod\limits_{k = {- \infty}}^{K}\; F} = {{\exp \left( {{{j2\pi} \cdot f_{T} \cdot K \cdot N_{Sym}} + \vartheta_{0}} \right)}.}}$

R_(K) is fed into a back rotation unit 520 where the complex conjugatesof these values are multiplied with the amplitude/phase estimator outputvalues

$P_{K} = {\sum\limits_{n = 0}^{N_{DFT} - 1}{{Y_{K}(n)} \cdot {{\overset{\sim}{W}}_{K}^{*}(n)}}}$

from an amplitude and phase estimation unit 570.Here {tilde over (W)}_(K)(n) is the estimation pattern, which equalsW(n) in a first embodiment of the invention, which may however besimplified in another embodiment, as explained below. Furthermore, theestimation pattern may vary from OFDM symbol to OFDM symbol, which isdenoted by the index K.

The obtained back-rotated amplitude/phase estimates Q_(K)=R_(K)·P*_(K)are fed into an Infinite Impulse Response (IIR) linear low-pass filter530 with a DC gain of one (“History averaging”), controlled by thefactors c_(K) with 0<c_(K)<1. In a first embodiment of the invention,the factors c_(K) are constant over time, irrespective of K.

The output values Q _(K) of filter 530 are then supplied to a forwardrotation unit 540 and are forward rotated to obtain the estimatedamplitude and phase for the current OFDM symbol, P _(K)= Q _(K)· R _(K).The filter output value Q _(K) is obtained after a filter memory,because the estimate applied for cancellation in the current OFDM symbolshould be based upon only previous OFDM symbols, thus exploitingstatistical independency.

The output value P _(K) of forward rotation unit 540 is then supplied toa pattern weighting unit 550 and is weighted by a cancellation patternŴ_(K)(n), which is, in a first embodiment of the invention, equal toW(n), but which may be simplified in another embodiment, as will beexplained below. Further, the cancellation pattern may vary from OFDMsymbol to OFDM symbol which is denoted by the index K. Finally, theobtained vector V_(K)(n) from pattern weighting unit 550 is fed into asubtractor 560 and is subtracted from vector Y_(K)(n) output by DFT unit420 of FIG. 4 to obtain an output vector after spur cancellation,

Z _(K)(n)=Y _(K)(n)−V _(K)(n).

In another embodiment of the invention the condition E{Y_(K)(n)}=0 maynot be satisfied for some pairs (K,n), which is the case if pilot tonesare included in the OFDM signal. To prevent the amplitude/phase estimatefrom becoming biased, the concerned pairs (K,n) shall not be consideredin the estimator.

This is achieved by a modified arrangement shown in FIG. 5 thatillustrates an extension of the spurious tone cancellation arrangementof FIG. 4. The components in FIG. 5 which are the same or equivalent tocomponents of FIG. 4 described above are designated with the samereference numerals, and a description thereof should not be repeated,for sake of brevity.

In the arrangement of FIG. 5, the output vectors from DFT unit 420 ofFIG. 3 are first fed into a pilot symbol replacement unit 580. In pilotsymbol replacement unit 580 all DFT output values, which shall not beconsidered in the estimation, are overwritten with the currentlyavailable estimate for the respective DFT bin. This is achieved bydefining a pattern S_(K)(n) which indicates at what positions the DFToutput values shall not be considered for the estimation. The values ofthis pattern are defined as 0 where the DFT output shall be consideredfor spur estimation, and otherwise as 1.

Thus, the functionality of pilot symbol replacement unit 580 may bedescribed by the equation

${U_{K}(n)} = \left\{ \begin{matrix}{Y_{K}(n)} & {\left. {\forall\left( {K,n} \right)} \middle| {S_{K}(n)} \right. = 0} \\{V_{K}(n)} & {{otherwise}.}\end{matrix} \right.$

Another embodiment of the invention exploits a degree of freedom in thechoice of the pattern W(n), which is to rotate the phase of the entirevector in the complex plane, in order to obtain real-valued coefficientsW(n), which reduces the computational complexity. This can also beachieved using the equation

${W(n)} = {\frac{1}{N_{DFT}} \cdot {\sum\limits_{k = 0}^{N_{DFT} - 1}{{\exp \left( {j\; 2{\pi \cdot \left( {{\left( {k - \frac{N_{DFT}}{2}} \right) \cdot f_{T}} - \frac{k \cdot n}{N_{DFT}}} \right)}} \right)}.}}}$

In still another embodiment of the invention the complexity of theamplitude/phase estimator 570 is reduced by exploiting the fact thatmost of the energy of the disturbing rotating exponential of knownfrequency is concentrated on a few bins at the DFT output. In thisembodiment only a subset of DFT output bins, indexed by the setN_(Est)={n₁, n₂, . . . , n_(N) _(Est) }⊂{0,1, . . . , N_(DFT)−1}

is used, and the estimation pattern is determined by

${{\overset{\sim}{W}}_{K}(n)} = \left\{ \begin{matrix}\frac{W_{K}(n)}{\sum\limits_{n \in N_{Est}}{{W_{K}(n)}}^{2}} & {\forall{n \in N_{Est}}} \\0 & {{otherwise}.}\end{matrix} \right.$

Here, the subscript K indicates that W(n) may vary from OFDM symbol toOFDM symbol. The set N_(Est) is typically defined such as to collectmost of the energy with a limited number of bins, which is achieved byusing only the coefficients with the largest absolute values in W(n). Inan extreme case, only a single value out of W(n) is used.

In still another embodiment of the invention the complexity of thepattern weighting/spur subtraction units, 550 and 560, respectively, isreduced by exploiting the fact that most of the energy of the disturbingrotating exponential of known frequency is concentrated on a few bins atthe DFT output, eliminating the need to subtract negligibly smalldisturbances. In this embodiment, only a subset of DFT output binsindexed by a set

N _(Cancel) ={n ₁ , n ₂, . . . , n _(N) _(Cancel) }⊂{0, 1, . . . , N_(DFT)−1}

is used, and the cancellation pattern is defined as

${{\hat{W}}_{K}(n)} = \left\{ \begin{matrix}{W_{K}(n)} & {\forall{n \in N_{Cancel}}} \\0 & {{otherwise}.}\end{matrix} \right.$

Again the subscript K indicates that W(n) may vary from OFDM symbol toOFDM symbol. The set N_(Cancel) is typically defined to apply to allelements in W(n) where an unacceptable excessive disturbance is expectedto occur. In an extreme case, only a single value out of W(n) isaddressed.

In another embodiment of the invention, a fast ring-in of the historyaveraging low-pass is realized by time-variation of the filtercoefficients c_(K). For example, when the first amplitude/phase estimateis performed at OFDM symbol with K=1, a good choice of a sequence is

$c_{K} = \left\{ \begin{matrix}0 & {K < 1} \\{1/K} & {1 \leq K < K_{Limit}} \\{1/K_{Limit}} & {K \geq {K_{Limit}.}}\end{matrix} \right.$

This results in an equal weighting of all incoming samples until thehistory averaging low-pass has rung in. After ring-in, weighting offiltered samples decays exponentially over time.

In another embodiment of the invention, each vector of samples subjectedto DFT is first cyclically shifted before the DFT is processed, due tothe OFDM receiver design. For a cyclic shift by N_(Shift) samples, theweighting pattern becomes

${W(n)} = {\frac{1}{N_{DFT}} \cdot {\sum\limits_{k = 0}^{N_{DFT} - 1}{{\exp \left( {j\; 2{\pi \cdot \begin{pmatrix}{\left( {\left( {k - N_{Shift}} \right){mod}\; N_{DFT}} \right) \cdot} \\{f_{T} - \frac{k \cdot n}{N_{DFT}}}\end{pmatrix}}} \right)}.}}}$

All other principles of the invention are applied as described before.

In another embodiment of the invention the frequency of the disturbingtone changes over time, possibly due to some adaptation of the mixerfrequency in the radio front-end. To cope with this, the offset phasorF_(K) as well as the estimation pattern {tilde over (W)}_(K)(n) and thecancellation pattern Ŵ_(K)(n) are adapted accordingly.

In another embodiment of the invention, where multiple disturbingsinusoids shall be cancelled, a plurality of spur cancellers, asdescribed above, may be implemented. In this case all amplitude/phaseestimations are performed in parallel on the DFT output data, whereasthe subtractions of the estimated tones occur sequentially, tone bytone.

As an example, consider a DVB-H receiver implementation withN_(DFT)=4096, N_(Guard)=1024, N_(Sym)=5120, N_(Shift)=512, with asampling frequency f_(sample)=48/7 MHz, which is disturbed by a spurioustone at a frequency f_(Spur)=1 MHz. The normalized frequency of the toneis f_(T)=f_(Spur)/f_(Sample)=7/48. The tone frequency corresponds withthe OFDM sub-carrier index n_(T)=f_(T)·N_(DFT)=597⅓. The offset phasoris determined as

$F_{K} = {{\exp \left( {j\; 2{\pi \cdot \frac{7}{48} \cdot 5120}} \right)} = {{- \frac{1}{2}} - {j{\frac{\sqrt{3}}{2}.}}}}$

The estimation pattern is defined as

${{\overset{\sim}{W}}_{K}(n)} = \left\{ \begin{matrix}\frac{2\pi}{3\sqrt{3}} & {{{for}\mspace{14mu} n} = 597} \\0 & {{otherwise},}\end{matrix} \right.$

the cancellation pattern is defined as

${{\hat{W}}_{K}(n)} = \left\{ \begin{matrix}\begin{matrix}{\frac{1}{4096} \cdot {\sum\limits_{k = 0}^{4095}{\exp \left( {j\; 2{\pi \cdot \begin{pmatrix}{\left( {\left( {k - 512} \right){mod}\; 4096} \right) \cdot} \\{f_{T} - \frac{k \cdot n}{4096}}\end{pmatrix}}} \right)}}} \\{{{for}\mspace{14mu} 591} \leq n \leq 603}\end{matrix} & \; \\0 & {{otherwise},}\end{matrix} \right.$

and the minimum filter constant after ring-in is set to

$C_{K,\min} = {\frac{1}{10}.}$

APPLICATIONS OF THE INVENTION

The various embodiments of the invention as detailed above may beapplied separately or in combination in an OFDM receiver for wireless orwired transmission including, but not limited to, receivers in wirelesslocal area network (WLAN) applications, e.g., according to the IEEE802.11 standard, in wireless personal area network (WPAN) applications,e.g., according to the IEEE 802.16 standard, in digital TV receiversfor, e.g., DVB-T, DVB-H, T-DMB, DMB-T, DAB, in ultra-wideband (UWB)receivers according to the multi-band OFDM alliance (MBOA) standardproposal, etc.

1. A method for cancelling additive sinusoidal disturbances in an OFDMreceiver, in which a digital complex baseband signal input is cut intoblocks of samples, the guard period thereof is removed, a DiscreteFourier Transform (DFT) is performed on a remainder of each sampleblock, and data equalization and symbol demapping is performed to outputsoft bits to be fed to a decoder, wherein a frequency of a disturbanceis known, the method being characterized by the steps of: generating atest signal representing a disturbing signal of a known frequency thatis presumed to occur in the OFDM receiver within a range of interest ofa RF receive signal; subjecting said test signal to DFT; estimating anamplitude and a phase of said disturbing signal using said test signalto obtain amplitude and phase estimation values (P_(K)); and eliminatingsaid disturbance from a received signal using the estimated amplitudeand phase.
 2. The method of claim 1 wherein said estimating stepcomprises obtaining amplitude and phase estimation values (P_(K)) byforming a scalar product of a current DFT output vector Y_(K)(n) with anestimation pattern ({tilde over (W)}_(K)(n)) which is only a function ofthe frequency of the disturbing signal.
 3. The method of claim 1 whereinsaid estimating step comprises obtaining said amplitude and phaseestimation values (P_(K)) by (i) forming a scalar product of a currentDFT output vector Y_(K)(n) with an estimation pattern ({tilde over(W)}_(K)(n)) which is only a function of the frequency of the disturbingsignal and (ii) overwriting at least one DFT output value which containsa pilot tone with a currently available estimate (V_(K)(n)) for therespective DFT bin.
 4. The method of claim 1 wherein only real-valuedcoefficients are considered for a weighting pattern.
 5. The method ofclaim 1 wherein only a subset of DFT output bins is considered for saidamplitude and phase estimation by using only coefficients with largestabsolute values for a weighting pattern.
 6. The method of claim 1wherein said estimation pattern may vary from OFDM symbol to OFDMsymbol.
 7. The method of claim 1 wherein the eliminating step comprisesthe steps of: cumulatively multiplying an offset phasor (F_(K)) betweentwo subsequent OFDM symbols derived from the DFT and obtaining asequence of start phasors, the start phasors having complex conjugates;multiplying the complex conjugates of said start phasors with saidamplitude and phase estimation values (P_(K)) to obtain back-rotatedamplitude/phase estimates (Q_(K)); obtaining an estimated amplitude andphase for a current OFDM symbol; weighting said estimated amplitude andphase by a cancellation pattern (Ŵ_(K)(n)) to obtain a vector(V_(K)(n)); subtracting the obtained vector (V_(K)(n)) from the DFToutput vector to achieve spur cancellation.
 8. The method of claim 7wherein only a subset of DFT output bins is considered for saidweighting and subtracting steps by using only coefficients with largestabsolute values for said cancellation pattern.
 9. The method of claim 7wherein said cancellation pattern may vary from OFDM symbol to OFDMsymbol.
 10. The method of claim 7 wherein the step of obtaining anestimated amplitude and phase for a current OFDM symbol compriseshistory averaging low-pass filtering said back-rotated amplitude/phaseestimates (Q_(K)) under control of a filter coefficient c_(K), with0<c_(K)<1.
 11. The method of claim 10 wherein said filter coefficientc_(K) is constant over time.
 12. The method of claim 10 wherein saidfilter coefficient c_(K) is variable over time.
 13. The method of claim1 wherein each vector of samples subjected to DFT is cyclically shiftedbefore the DFT process.
 14. The method of claim 7 wherein, if thefrequency of the disturbing signal changes over time said offset phasor(F_(K)), said estimation pattern ({tilde over (W)}_(K)(n)) and saidcancellation pattern (Ŵ_(K)(n)) are adapted accordingly.
 15. The methodof claim 1, wherein if the method is effected for a plurality ofdisturbance signals of different frequencies, all amplitude/phaseestimations are performed in parallel on the DFT output data and thesubtractions of the estimated signals are performed sequentially, signalby signal.
 16. An arrangement for cancelling additive sinusoidaldisturbances in OFDM receivers, in which a digital complex basebandsignal input is cut into blocks of samples, a guard period thereof isremoved, a Discrete Fourier Transform (DFT) is performed on a remainderof each sample block, and data equalization and symbol demapping isperformed to output soft bits to be fed to a decoder, wherein thefrequency of a disturbance is known, the arrangement comprising: meansadapted for cumulatively multiplying an offset phasor (F_(K)) betweentwo subsequent OFDM symbols derived from the DFT and for obtaining asequence of start phasors, the start phasors having complex conjugates;amplitude and phase estimation means adapted for obtaining amplitude andphase estimation values (P_(K)) by forming a scalar product of a currentDFT output vector Y_(K)(n) with an estimation pattern ({tilde over(W)}_(K)(n)) which is only a function of the frequency of the disturbingtone; means adapted for multiplying the complex conjugates of said startphasors with the values (P_(K)) supplied by said amplitude and phaseestimation means to obtain back-rotated amplitude/phase estimates(Q_(K)); means adapted for obtaining an estimated amplitude and phasefor a current OFDM symbol; means adapted for weighting said estimatedamplitude and phase by a cancellation pattern (Ŵ_(K)(n)) to obtain avector (V_(K)(n)); and means adapted for subtracting the obtained vector(V_(K)(n)) from the DFT output vector to achieve spur cancellation. 17.The arrangement of claim 16, further comprising overwriting meansadapted for overwriting at least one DFT output value which contains apilot tone with a currently available estimate (V_(K)(n)) for therespective DFT bin.